AN1526 Freescale Semiconductor / Motorola, AN1526 Datasheet - Page 14

• Manufacturer Part Number: AN1526
• Description: RF Power Device Impedances: Practical Considerations
• Manufacturer: Freescale Semiconductor / Motorola

CONCLUSIONS
the input and output large–signal device impedances are not
only frequency dependent, but are also determined by the
operating conditions of the device. Because of the wide
range of possible applications, it is virtually impossible for
the device manufacturers to present impedance data for
every eventuality. The user, therefore, is left with the choice
of either measuring the device impedances under the
conditions he plans to use the device, or resorting to the
classical methods of tweaking the circuit impedances into
approximately the optimum match. The future does hold
some promise in two areas. Automated tuners will enable
impedance data to be gathered faster, enabling more
comprehensive data to be included on the data sheets with
the eventual possibility of publishing device impedance
distributions. Compact device models in conjunction with
non–linear simulators hold the best hope for simulating the
device under the proposed operating conditions, and then
permitting the software to synthesize the optimum broadband
matching networks.
equipment and methods. For the load–pull measurements
described earlier in this paper we used readily available and
inexpensive equipment. In addition to the usual equipment
found on an RF power bench including a computer as the
instrument controller, the only additional pieces of equipment
needed are a vector voltmeter, a variety of low attenuation
power attenuators, and a variable length shorted stub.
Figure 24 shows a block diagram of the bench set–up. A
series of load mismatch conditions was established by
terminating a broadband test fixture with the attenuators and
shorted stub. The shorted stub was calibrated at
approximately 20 intervals to establish varying phase shifts.
By varying the value of attenuation, a grid of load
impedances can be presented to the device on a network
of VSWR circles in the reflection coefficient plane. The
system was first calibrated by using a network analyzer and
a probe in the device socket to measure this series of load
impedances. A vector voltmeter, with error correction, could
of course have been used to measure these impedances.
the drive level adjusted to obtain rated output power under
optimum tuning for maximum gain into a matched load. With
the drive level fixed at this level, the output power was
remeasured over the range of calibrated load impedances.
This procedure was repeated at each frequency desired. The
input match was tuned for zero reflected power with the
output terminated in a matched load. The input return loss,
under mismatched conditions, thereby indicates changes in
magnitude of the input impedance. In addition to the usual
14
SuperCompact is a trademark of Compact Software and
Touchstone is a registered trademark of EESof, Inc.
For an RF power transistor we have demonstrated that
Load–pull measurements can employ a variety of test
After system calibration, the transistor was operated with
Load–Pull Method and Corrections for
Non–50 Ohm Environment
Power Measurement in
APPENDIX I:
Freescale Semiconductor, Inc.
For More Information On This Product,
Go to: www.freescale.com
MOTOROLA SEMICONDUCTOR APPLICATION INFORMATION
power meter to measure the output power from the test
fixture, a vector voltmeter monitors the test fixture load
reflection coefficient
was applied to the measured reflection co–efficient and this
value is then used to correct the output power, P
system is calibrated over a range of frequencies and the error
correction in software.
reading, [16]
where
and
hence efficiency can be obtained for a system in which the
load impedance is perturbed from the characteristic
impedance of the transmission line power meter
components. Contours can be generated from this grid data
by a number of commercially available software packages.
of the attenuator/shorted stub combination. The tuners can
be either manual or automated. The advantage of the latter
is that with suitable software the de–embedded load
i mpe dan c e pres ent ed to the d ev ic e is a v ai labl e
instantaneously. Also, with suitable software, the gain and
efficiency circles can be determined by contour following
techniques in real time, instead of fitting contours to
measurements on a grid of load mismatch points [20].
REFERENCES
(1) K. Kurokawa, “Power Waves And Scattering Matrix,”
(2) R. J. Frost, “Large–Signal S–Parameters Help Analyze
(3) R. Hejhall, “Small–Signal Parameters Aid FET
(4) R. J. Chaffin, and W. H. Leighton, “Large–Signal
(5) L. S. Houselander, H. Y. Chow, and R. Spence,
(6) O. Müller, “Large–Signal S–Parameter Measurements
The following formula can be used to correct the power
Using this method, accurate measurement of power and
An alternative system would be to use tuners in place
IEEE
Techniques, 194–202, March 1965.
Stability,” Electronic Design, 93–98, May 24, 1980.
Power–Amp Design,” Microwaves & RF, 141–144,
September, 1985.
S–Parameter
Transistors,” Digest of Technical Papers, Proceedings
1973 IEEE MTT International Microwave Symposium,
University of Colorado, Boulder, June 5, 1973.
“Transistor Characterization by Effective Large–Signal
2–Port Parameters,” IEEE Journal of Solid–State
Circuits, SC–5, 77–79, April 1970.
of Class C Operated Transistors,” Nachrichtentech Z.,
October 1968.
e
and frequency response errors determined by
normal vector analyzer correction techniques [32],
P
for directional coupler coupling magnitude,
attenuation and meter frequency response.
23
M
om
, e
is the uncorrected load reflection coefficient,
P
is the measured power meter reading corrected
out
L
21
Transactions
= e
, and e
= P
23
om
+
L
Characterization
22
1 – e
. Standard three term error correction
e
22
e
1 – e
e
24
24 M
21 M
are the directivity, source match
on
21
M
Microwave
2
(1 –
of
L
UHF
2
Theory
)
out
Power
. The
and
(2)
(3)